Resonant frequency tracking system and method for use in a radio frequency (RF) power supply

ABSTRACT

A radio frequency (RF) power system for providing RF power to a load. The RF power system comprises a tank circuit; a direct current (DC) voltage source that provides a DC voltage within a first predetermined range; an amplifier, coupled to said DC voltage source, to provide an alternating voltage to said tank circuit; a frequency controller, coupled to said amplifier, to control a frequency of said alternating voltage provided by said amplifier; and a power sensor being coupled to said tank circuit for providing a signal to said frequency controller, wherein said frequency controller controls said frequency of said alternating voltage based on said signal provided from said power sensor.

The present invention is a divisional of U.S. patent application Ser.No. 09/113,518 entitled “RF Power Supply” by Thompson et al., assignedto the present assignee and filed on Jul. 10, 1998.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to the field of radio frequency(RF) power supplies. The present invention is useful in inductionheating and plasma applications, but by no means is limited to suchapplications.

2. Discussion of the Background

Induction heating is a method of heating electrically conductivematerials such as metals. Induction heating relies on, as the nameimplies, inducing electrical currents within a material to be heated.These induced currents, called eddy currents, dissipate energy and bringabout heating. Common uses of induction heating include heat treating,welding, melting, packaging and curing. The number of consumer itemswhich undergo induction heating during some stage of their production islarge and rapidly expanding.

Prior to the development of induction heating, gas and oil-firedfurnaces provided the prime means of heating metals and nonmetals. Theadvantages that induction heating offers over furnace techniques arenumerous. For example, greater heating rates can be achieved byinduction heating than can be achieved by gas or oil furnaces. Higherheating rates lead to shorter heating times, which lead to productivityincreases and reduced labor costs. Furthermore, given today'senvironmental concerns, induction heating is an attractive alternativeto pollution producing furnaces.

The basic components of an induction heating system are (1) an AC powersource (RF) power supply, (2) a tank circuit having an inductor coil anda capacitor, and (3) the material to be heated (a.k.a., “workpiece” or“load”). Common tank circuits used in induction heating are eitherparallel resonant or series resonant. A parallel resonant tank circuitincludes a capacitance in parallel with the inductor coil and a seriesresonant tank circuit includes a capacitance in series with the inductorcoil. A workpiece is heated by placing the workpiece within the inductorcoil of the tank circuit and applying a high-power, RF alternatingvoltage to the tank circuit using the power supply. (The alternatingvoltage applied to the tank circuit causes an alternating current toflow through the inductor coil. The flow of an alternating currentthrough the inductor coil generates an alternating magnetic field thatcuts through the workpiece placed in the inductor coil. It is thisalternating magnetic field that induces the eddy currents that heat theworkpiece.)

A workpiece is heated most efficiently when the frequency of thealternating voltage applied to the tank circuit matches the tankcircuit's resonant frequency. That is, when the tank circuit (i.e., thetank circuit with a workpiece placed in the inductor coil) is driven atits resonant frequency, the transfer of power from the power supply tothe workpiece is maximized. Thus, heating of the workpiece at theresonant frequency yields the greatest heating efficiency.

It should be noted that the resonant frequency of the tank circuit is inpart determined by the characteristics of the inductor coil, such as thesize and shape of the coil, and the characteristics of the workpiecewhen the workpiece is placed in the coil. Hence, moving the workpiecethrough the coil or altering the characteristics of the workpiece byheating it will change the resonant frequency of the tank circuit.Because the resonant frequency of the tank circuit changes as theworkpiece is heated or moved through the coil, induction heating systemsutilize a power supply having a tuning system for continuously trackingthe resonant frequency of the tank circuit. By tracking the resonantfrequency of the tank circuit, the power supply is better able toprovide an alternating voltage that matches the resonant frequency,thereby efficiently heating the workpiece.

A problem with conventional induction power supplies, however, is thatthey operate over a limited frequency band. Another problem is that theyare not capable of delivering a power into a load that is remotelylocated from the power supply. Therefore, what is desired is an RF powersupply that overcomes the above and other limitations of conventional RFpower supplies.

SUMMARY OF THE INVENTION

The present invention provides an RF power supply that is capable ofquickly responding to varying load conditions so as to deliver thedesired amount of power to the load. The RF power supply according tothe present invention can track rapid changes in the resonant frequencyof a tank circuit. The present invention also provides an RF powersupply capable of delivering a wide range of power over a broadfrequency range to a load that is remotely located from the powersupply. The ability to deliver a wide range of power over a broadfrequency range is a significant advantage because it enables anoperator of the RF power supply to efficiently heat a wide variety ofwork pieces without having to change any components of the RF powersupply.

According to one embodiment, the RF power supply includes a DC voltagesource that provides a DC voltage within a predetermined voltage range;an amplifier, coupled to the DC voltage source, that provides analternating voltage to a circuit connected to the RF output of the RFpower supply; a frequency controller, coupled to the amplifier, to setthe frequency of the alternating voltage produced by the amplifier; anda sensor, coupled to the circuit, to provide a signal to the frequencycontroller, where the frequency controller sets the frequency of thealternating voltage based on the signal received from the sensor. In oneembodiment, the circuit is a tank circuit.

In one embodiment, the voltage source receives an AC voltage andconverts it to a DC voltage. Preferably, the DC voltage source includesa pulse width modulator with hysteretic current mode control. Theadvantage of using such a pulse width modulator is that the DC voltagethat is provided to the amplifier remains constant regardless ofvariations in the load and regardless of changing AC line voltages orfrequencies. This is advantageous because the desired power level willbe delivered to the load even when the load varies, regardless ofchanging AC line voltages or frequencies. Another significant advantageis that the DC voltage source according to a preferred embodiment isable to rapidly vary its output voltage over a wide range, therebyproviding a means for rapidly varying the power delivered to the loadover a wide power range.

In one embodiment, the frequency controller includes a processor and afrequency synthesizer. The processor receives a sensor signal from thesensor and, based on the received sensor signal, sends a frequencycontrol signal to the frequency synthesizer. The frequency synthesizeroutputs an alternating voltage, where the frequency of the alternatingvoltage is controlled by the frequency control signal. The output of thefrequency synthesizer is coupled to the amplifier. The amplifierproduces an alternating voltage having the same frequency as thefrequency of the signal outputted by the frequency synthesizer. In thismanner, the frequency controller sets the frequency of the alternatingvoltage produced by the amplifier. The advantage of using a processorand frequency synthesizer to set the frequency of the alternatingvoltage produced by the amplifier is that it enables the RF power supplyto (1) quickly respond to varying load conditions; (2) operate over awide range of frequencies; (3) easily adapt to series and parallelresonant tank configurations; and (4) easily adapt to a wide variety ofresonant sensing schemes, such as an admittance, an impedance, acurrent, or a reflected power resonant sensing scheme.

In one embodiment, the sensor is an admittance sensor and the sensorsignal provided to the frequency controller represents an admittance ofthe tank circuit. The admittance sensor provides numerous advantages.For example, (1) the admittance sensor enables the RF power supply totrack the resonant frequency during rapid voltage ramp periods and overa broad frequency range, and (2) because the admittance sensor istolerant of various waveshapes encountered in RF transmissions, theadmittance sensor can be located at the tank circuit, thereby allowingremote sensing. In another embodiment, the sensor is a forward and/orreflected power sensor, and the signal provided to the frequencycontroller represents the forward power, the reflected power, the ratioof the forward to reflected power, or the ratio of the reflected toforward power.

The present invention additionally provides a unique method fordelivering RF power to a load. The method quickly determines theresonant frequency of the tank circuit and is able to track rapidchanges in the resonant frequency. These and further features andadvantages of the present invention, as well as the structure andoperation of various embodiments of the present invention, are describedin detail below with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated herein and form partof the specification, illustrate the present invention and, togetherwith the description, further serve to explain the principles of theinvention and to enable a person skilled in the pertinent art to makeand use the invention. In the drawings, like reference numbers indicateidentical or functionally similar elements. Additionally, the left-mostdigit(s) of a reference number identifies the drawing in which thereference number first appears.

FIG. 1A illustrates an RF power supply according to one embodiment.

FIGS. 1B and 1C illustrate a parallel resonant tank circuit and a seriesresonant tank circuit, respectively.

FIG. 2 illustrates a frequency controller according to one embodiment.

FIG. 3A illustrates a sensor according to a first embodiment.

FIG. 3B illustrates a sensor according to a second embodiment.

FIG. 4A illustrates a DC voltage source according to one embodiment.

FIG. 4B illustrates a snubber circuit according to one embodiment.

FIG. 5 is a flow chart illustrating a process for inductively heating aworkpiece.

FIG. 6A illustrates a course tuning process according to one embodiment.

FIG. 6B illustrates a fine tuning process according to one embodiment.

FIG. 7A illustrates a resonant frequency tracking process according to afirst embodiment.

FIG. 7B illustrates a resonant frequency tracking process according to asecond embodiment.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1A is a block diagram illustrating the components of an RF powersupply 100 according to one embodiment of the present invention. RFpower supply 100 includes a direct current (DC) voltage source 102 forproviding a controlled DC voltage to amplifier 104. Amplifier 104receives a low voltage RF signal from frequency controller 112 andamplifies that RF signal to provide an alternating voltage to tankcircuit 108. The power delivered to load 109 (see FIG. 1B) is determinedby the frequency of the alternating voltage and the level of the DCvoltage provided to amplifier 104.

In one embodiment, the DC voltage produced by DC voltage source 102ranges between 0 and 350 volts, and the frequency range of frequencycontroller 112 is 1 KHz to 15 MHz. Consequently, RF power supply 100 isable to deliver a wide range of power over a wide frequency range.Furthermore, RF power supply 100 is able to deliver the wide range ofpower over the wide frequency range over long distance RF cables 107,thereby allowing tank circuit 108 to be remotely located from powersupply 100. Additionally, frequency controller 112 and sensor 114 enableRF power supply 100 to track rapid changes in the resonant frequency andquickly respond to varying load 109 conditions so as to continuouslydeliver the desired amount of power to load 109. An embodiment of DCvoltage source 102, amplifier 104, frequency controller 112, and sensor114 will be described in more detail below with reference to theaccompanying drawings. DC Voltage source 102.

In one embodiment, DC voltage source 102 receives an alternating current(AC) voltage from an AC power source (not shown). The DC voltage source102 converts the AC voltage provided by the AC power source to a DCvoltage. In one embodiment, the DC voltage produced by DC voltage source102 ranges between 0 and 350 volts. A main controller 110 is coupled toDC voltage source 102 and a control panel 116. An operator of RF powersupply 100 sets a desired voltage level via the control panel 116. Themain controller 110 controls the voltage level of the DC voltageproduced by the DC voltage source 102 according to the voltage level setby the operator. In a preferred embodiment, DC voltage source 102 iscapable of maintaining a constant DC output voltage, current, or power(depending on the application's requirements) regardless of changes inload 109.

FIG. 4A illustrates a preferred embodiment of DC voltage source 102. Thedesign of DC voltage source 102 must take a variety of factors intoconsideration. For example, DC voltage source 102 must be capable ofproviding maximum current at any set output voltage from approximately5% to 100% of maximum output voltage and it must quickly compensate forchanging load conditions. In one embodiment, the maximum output voltageis 350 Volts. Additionally, DC voltage source 102 must be capable oframping voltage up and down rapidly to provide controlled heat cycleswhen performing pulse/heating applications. Furthermore, the ability totightly regulate the output of DC voltage source 102 in a constantvoltage, constant current, or constant power application is also highlydesirable. High efficiency and power factor are also desirable goals.

The DC voltage source 102 illustrated in FIG. 4A is designed to meet theabove requirements. The preferred embodiment of DC voltage source 102illustrated in FIG. 4A is an off-line current mode buck regulatortopology pulse width modulator. This configuration provides inherentshort circuit protection and input ripple rejection, has simple loopstability design, and has instantaneous response to changes in loadcurrent.

Soft start circuit 402 is included. Soft start circuit 402 limits inrushcurrents to a reasonable level. In addition to this basic functionality,missing phase detection is performed to prevent the supply from turningon in the event of a missing phase. Conversion of the AC input to DC isperformed in rectifier/filter circuitry 404. The rectified and filteredinput voltage is then presented to the pulse width modulator (PWM) 410.

PWM 410 includes a Snubber 412, a Switch 414, a Controller 416, and anError Amplifier 418. PWM 410 hard switches its DC input voltage at acontrolled pulse width and duty cycle. These pulses of energy are thenfiltered through an inductor 420 and capacitor 422 to produce thedesired output voltage 490. Preferably, inductor 420 has an inductanceof 1.8 mili-Henries (mH) and capacitor 422 has a capacitance of 390micro-Farads (uF) when the input voltage is 480 VAC.

A free-wheeling diode 424 is connected to switch 414 output to enablecurrent to continue to flow through inductor 420 when switch 414 is off.To maximize efficiency and simplify drive requirements for the powerlevels provided by DC voltage source 102, an Insulated Gate BipolarTransistor (IGBT) 470 (see FIG. 4B) was chosen for the switch 414. Thisdevice provides the turn-on characteristics of MOSFETS, and the superiorsaturation characteristics of a bipolar transistor.

Depending on the input voltage version of the supply, the basic designhas been applied to power supplies providing up to 70 kW at voltages to350 VDC and currents to 200 amperes.

FIG. 4B illustrates snubber circuit 412 according to one embodiment. Thesnubber circuit 412 is designed to provide zero current switching at theturn-on of switch 414, thereby minimizing the tun-on transition powerdissipation normally associated with buck regulators due to the rate ofvoltage change, the reverse recovery current of the free-wheeling diode424, and the current due to the load. Functionally, snubber circuit 412shown in FIG. 4B delays the reverse recovery current of the freewheeling diode 424 from appearing on the switch 414 until the switch 414is fully on. This prevents excessive power loss from occurring in theswitch 414 during the ON time transition, as the recovery current isdelayed until the switch 414 is fully on.

Snubber circuit 412, according to one embodiment, uses an inductor 484to soften and delay the freewheeling diode 424 reverse recovery current,and a resonant LC snubber (481 and 485) to prevent voltage spikes on theswitch output (emitter) 472 which would be caused by use of the inductoralone.

Starting with the switch 414 in the ON state, the followingvoltage/current conditions are present: the switch emitter 472 voltageis equal to Vin, which is the switch input (collector) 474 voltage. Thecurrent through inductor 484 is at maximum. Free-wheeling diode 424cathode voltage is equal to switch emitter 472 voltage and the currentthrough it is zero. The current through inductor 481 is zero. Thevoltage of capacitor 485 is at its negative peak, referenced to theswitch emitter 472, and its current is zero. The capacitor 485 voltageis proportional to the current through inductor 484, based on equalenergy, C V²=LI².

When the switch 414 turns off, the current through switch emitter 472falls to zero, and load current now circulates through inductor 484 viadiode 480 and capacitor 485. Emitter 472 voltage transitions towardszero rapidly, until its voltage is equal to the peak voltage of thecapacitor 485. At this point, the voltage starts decaying at a constantrate equal to the capacitor 485 current (which equals the constantcurrent through inductor 484) divided by the capacitance (E/T=I/C). Thecapacitor 485 is charging from its negative peak towards its positivepeak, starting from the initial emitter 472 turn-off. When the emitter472 voltage crosses zero, the free-wheeling diode 424 turns ON, currentthrough inductor 484 starts falling towards zero, the voltage acrosscapacitor 485 is zero, and current in the free-wheeling diode 424 startsincreasing towards the full load current. The capacitor 485 is nowcharged in a resonant mode by the energy in inductor 484 until itspositive peak is reached.

At switch 414 turn-on the emitter 472 voltage rises rapidly to Vin andthe capacitor 485 voltage starts decaying resonantly. In a classic buckregulator configuration, the free-wheeling diode 424 is connecteddirectly to the switch emitter 472, and is turned on rapidly at theswitch 414 turn-on point. Thus any diode reverse recovery current isseen immediately by the switch 414. In the snubber configurationaccording to one embodiment, diode 424 turn-on is delayed after switch414 turn-on so that any reverse recovery current is not seen by theswitch 414 until it is saturated. As an added benefit, diode 424 reverserecovery is softened by controlling the rate of change of turn offcurrent (di/dt) through inductor 484. This is accomplished as follows:when the switch 414 turns on, current starts flowing through the switch414 and inductor 484 at a di/dt rate of Vin/L, displacing thefree-wheeling diode 424 current until maximum current is flowing throughthe inductor 484 and the diode 424 current is zero. Due to stored chargein the diode 424, negative current flows in the diode 424 until thestored charge is dissipated. When the diode 424 current reaches itsnegative peak, the diode 424 turns off, and its reverse recovery currenttransitions to the switch 414 at a time t=(LI/E) after switch turn-on.The energy stored in the diode 424 from the zero crossing to thenegative peak is dissipated in the switch 414, with the remaining energydissipated in the diode 424. Energy stored in the capacitor istransferred to inductor 481 and then back to the capacitor via ahalf-sinusoidal current pulse (resonant transfer at the frequency of thetank formed by capacitor 485 and inductor 481), circulating through theswitch. Diode 482 blocks Vin from creating a DC offset in the snubberand diode 483 clamps transient voltage spikes occurring on afree-wheeling diode 424 to Vin.

Referring again to FIG. 4A, controller 416 provides the gating signal447 for the switch 414. Preferably, controller 416 is a hystereticcurrent mode PWM integrated circuit. This device uses current feedbackfrom the output inductor 420 to determine the operating frequency anduses voltage feedback to determine the output voltage.

When DC voltage source 102 is operating in a continuous mode, the acripple current in the output inductor 420 is triangular in shape. Acurrent transducer 426 samples this ac ripple current, as well as the DCcurrent in the inductor 420, and feeds a current feedback signal 443 tothe controller IC 416. The desired peak-to-peak ripple current (Delta I)is maintained at a constant level by the hysteretic controller 416,which adjusts the switching pulse width and switching frequency ofswitch 414 to maintain Delta I at its predetermined value while alsomaintaining the preset output voltage 490.

Error amplifier 418 compares the desired output voltage level, which itreceives from main controller 110 through communications interface 441,to an output voltage feedback signal 442 and generates an error signal444 when a differential is present. When the output voltage 490 is lowerthan the desired output voltage level, the error signal 444 that isgenerated by error amplifier 418 will cause controller 416 to turnswitch 414 ON. Conversely, when the output voltage 490 is higher thanthe desired output voltage level, the error signal 444 that is generatedby error amplifier 418 will cause controller 416 to turn switch 414 OFF.In this manner the desired output voltage will be maintained.

Error amplifier 418, in conjunction with the Fault Detection andProtection block 430, also provides a current limiting function. When anovercurrent is detected by Fault Detection and Protection block 430, anovercurrent signal 445 is sent to the error amplifier 418, which adjuststhe output voltage via error signal 444 to maintain the preset currentlimit.

Fault detection and protection block 430 continuously samples the inputvoltage, output voltage feedback signal 442, and output current feedbacksignal 443. If any of these signals go out of established limits, afault signal 446 is generated and sent to the main controller 110 viacommunications interface 441. Upon receiving the fault signal 446, maincontroller 110 turns the system off.

Amplifier 104

Amplifier 104 is coupled to DC voltage source 102 and frequencycontroller 112. Amplifier 104 is also coupled to tank circuit 108through transformer 106. The function of transformer 106 is to provideload isolation. Amplifier 104 provides an alternating voltage to tankcircuit 108. The frequency of the alternating voltage provided to tankcircuit 108 is controlled by frequency controller 112. The peak voltagelevel of the alternating voltage is determined by the output voltage 490provided by DC voltage source 102. Amplifier 104, in one embodiment,includes a switch mode amplifier. Preferably, amplifier 104 includes afull-bridge switch mode amplifier having an inductive clamp topology,which is capable of recovering all reactive energy. Such a full-bridgeswitch mode amplifier is described in copending U.S. patent applicationNo. 09/113,522, entitled “System For Enabling a Full-Bridge Switch-ModeAmplifier to Recover All Reactive Energy,” filed by Dan Lincoln,assigned to the assignee of the present invention and incorporated byreference in its entirety herein.

In a preferred embodiment, amplifier 104 is able to deliver analternating voltage having a frequency between 1 KHz and 15 MHz to tankcircuit 108. One skilled in the relevant art will recognize that thereare a variety of amplifiers for producing alternating voltages, and thatthe present invention is not limited to any particular amplifier.

Frequency Controller 112

Frequency controller 112 is coupled to amplifier 104 and to tank circuit108 through sensor 114. The function of frequency controller 112 is totrack the resonant frequency of tank circuit 108 based on the output ofsensor 114 and to control the frequency of the alternating voltageproduced by amplifier 104 such that the frequency of the alternatingvoltage matches the resonant frequency of the tank circuit 108. Byperforming this function, the desired amount of power will be deliveredto load 109 regardless of variations in load 109 that develop as poweris delivered to load 109.

Frequency controller 112 tracks the resonant frequency by monitoring theoutput of sensor 114. Preferably, sensor 114 is an admittance sensor andthe signal fed into frequency controller 112 from sensor 114 representsthe admittance of tank circuit 108. In another embodiment, sensor 114senses reflected and/or forward power. Both embodiments of sensor 114are described in more detail in another section of this document.

The tank circuit 108 may be either a parallel resonant or seriesresonant tank circuit in combination with a workpiece 109. FIGS. 1B and1C illustrate a parallel resonant tank circuit 108 b and a seriesresonant tank circuit 108 c, respectively. When the parallel resonanttank circuit 108 b is driven at its resonant frequency the currentflowing into the circuit is at a minimum and the voltage seen by thetank circuit 108 b is at a maximum. Admittance (Y) is defined as current(I) divided by voltage (V) (Y=I/V). Thus, when the parallel resonanttank circuit 108 b is driven at its resonant frequency, the tank circuit108 b has a minimum admittance. Consequently, the resonant frequency ofparallel resonant tank circuit 108 b can be determined by the frequencythat produces the minimum admittance.

For the case where tank circuit 108 is a series resonant tank circuit108 c and the series tank circuit 108 c is driven at its resonantfrequency, the current flowing into the tank circuit 108 c is a maximumand the voltage is a minimum (hence, the admittance is a maximum). Thus,for the series resonant tank circuit 108 c, the resonant frequency canbe determined by the frequency that produces the maximum admittancevalue.

FIG. 2 illustrates one embodiment of frequency controller 112. As shownin FIG. 2, one embodiment of frequency controller 112 includes: aprocessor 202 for tracking the resonant frequency of tank circuit 108and controlling the frequency of the alternating voltage; a frequencysynthesizer 204 for generating a signal having a frequency between 0 and20 MHz; a digital delay device 206; a logic device 208; and a driver210. Preferably, frequency synthesizer 204 is a direct digitalsynthesizer (DDS).

Based on signals received from sensor 114, processor 202 determines andtracks the resonant frequency of tank circuit 108 and directs frequencysynthesizer 204 to output a signal having a frequency matching theresonant frequency. FIGS. 7A and 7B each illustrate a procedure that canbe performed by processor 202 for tracking the resonant frequency (theseprocedures will be described in more detail below). The output offrequency synthesizer 204 is coupled to amplifier 104 through delaycircuit 206, logic circuit 208, and driver 210. Digital delay device 206and logic device 208 are used to generate two gating signals that are180 degrees out of phase with respect to each other. The frequency ofthe gating signals outputted from logic device 208 matches the frequencyof the signal fed to digital delay circuit 206 from frequencysynthesizer 204. The output of logic device 208 is fed to a drivercircuit 210 that provides the gating signals to the switches withinamplifier 104. Amplifier 104 produces an alternating voltage having afrequency that matches the frequency of the signal produced by thefrequency synthesizer 204. In this manner, the frequency controllerensures that the amplifier will drive the tank circuit 108 at the load'sresonant frequency. In one embodiment, logic device 208 is a complexprogrammable logic device (CPLD). In another embodiment, logic device isa processor controlled by the appropriate software. An advantage of thefrequency controller according to the preferred embodiment is that it isable to track rapid changes (e.g., 2 KHz/millisecond) in the resonantfrequency of tank circuit 108. Sensor 114.

As stated above, Preferably, sensor 114 is an admittance sensor thatprovides a signal representative of the admittance of tank circuit 108to processor 202. In other embodiments, sensor 114 could be a current,voltage, phase, impedance, or forward/reflected power sensor.

FIG. 3A illustrates a preferred embodiment of sensor 114. A preferredembodiment of sensor 114 includes a current sensor 310 and a voltagesensor 320 for measuring the current flowing into tank circuit 108 andthe voltage seen by tank circuit 108, respectively, so that theadmittance of tank circuit 108 can be determined. A preferred embodimentof sensor 114 further includes analog divider 330 to receive a signalfrom current sensor 310 and voltage sensor 320 and to output a signalrepresenting the admittance of tank circuit 108. This admittance signalis converted to a digital signal by A/D converter 340 and fed intofrequency controller 112.

In one embodiment, current sensor 310 includes a full wave bridge 312, afilter 314, and a current gain stage 316. Similarly, voltage sensor 320includes a full wave bridge 322, a filter 324, and voltage gain stage326. Full wave bridges 312, 322 produce a rectified current signal andvoltage signal, respectively. The rectified current signal is filteredto produce an average current signal and the rectified voltage signal isalso filtered to produce an average voltage signal. The average currentand voltage signals are then scaled by current gain stage 316 andvoltage gain stage 326, respectively. The current gain stage 316 andvoltage gain stage 326 are set such that the output of analog divider330 is approximately 40% of the maximum allowable voltage for Analog toDigital (A/D) converter 340 when the admittance of tank circuit 108 isequal to a nominal admittance. The nominal admittance is based on themaximum power that can be delivered to load 109 when DC voltage source102 is operating at its maximum output voltage of 350 volts. The outputof analog divider 330 was optimized to this nominal admittance to give a100 to 1 dynamic admittance range, thus allowing frequency controller112 to find the resonant frequency for tank circuit 108 having a qualityfactor (Q) ranging from 3 to 200. By changing either the voltage gainstage 326 or current gain stage 316, frequency controller 112 will beable to operate with either higher or lower Qs. Preferably, currentsensor 310 and voltage sensor 320 are placed at the tank circuit 108.That is, current sensor 310 and voltage sensor 320 are connected at thefar end of RF cables 107, as is shown in FIG. 3, so that the effects ofRF cable 107 are minimized, thereby assuring an accurate admittancereading of tank circuit 108. RF cables 107 are used to remotely locatethe tank circuit 108 from the power supply. In one embodiment, RF cablescan have a length of 200 feet without significantly effecting thedelivery of power from the power supply 100 to the load 109.

FIG. 3B illustrates an alternative embodiment of sensor 114. In thealternative embodiment, sensor 114 is a reflected and forward powersensor. As shown in FIG. 3B, sensor 114, according to the alternativeembodiment, includes a first transformer 350, a second transformer 352,a first resistor 353, a second resistor 354, a first peak detect circuit380, and a second peak detect circuit 382. The first peak detect circuit380 includes a first diode 356 and a first capacitor 362 connected inseries. The second peak detect circuit 382 includes a second diode 358and a second capacitor 364 connected in series. The voltage across thefirst capacitor 362 represents the forward power and the voltage acrossthe second capacitor represents the reflected power. The voltage acrossthe first capacitor 362 and the voltage across the second capacitor 364are fed into analog divider 330 by connections 372 and 370,respectively. The output of analog divider 330 is a signal representingthe ratio of forward power to reflected power.

In one embodiment, the first transformer 350 has a primary to secondaryturns ratio of X:1, and the second transformer 352 has a primary tosecondary turns ration of 1:Y. The optimal resistance (R) for the firstresistor 353 and the second resistor 354 is given by the followingformula: R=Y×R_(t)/X, where R_(t) is the resistance of tank circuit 108.

When the frequency of the alternating voltage provided to the tankcircuit 108 matches the tank circuit's resonant frequency, the reflectedpower is at a minimum point, the forward power is at a maximum point,and the ratio of reflected power to forward power is at a minimum point.This is true whether the tank circuit 108 is a parallel resonant tankcircuit 108 b or a series resonant tank circuit 108 c. Consequently, thefrequency controller 112 can use the ratio of the reflected power to theforward power, the reflected power, or the forward power to track thetank circuit's resonant frequency.

Process for Heating a Workpiece Through Induction

FIG. 5 is a flow chart illustrating a process for inductively heating aworkpiece 109 placed within an inductor coil 113 of a tank circuit 108using RF power supply 100. The process begins with step 502 when a user117 activates a “heat-on” button (not shown) on the control panel 116,which sends a “heat-on” signal to the main controller 110. Uponreceiving the “heat-on” signal, the main controller 110 begins theinitial tuning process for determining a precise or “fine” estimate ofthe tank circuit's resonant frequency. The initial tuning processencompasses steps 504-508. In step 504, main controller 110 commands DCvoltage source 102 to output a “tune” voltage. The “tune” voltage is thelowest voltage level that can provide a sufficient signal to measure theadmittance, impedance, reflected power, or forward power of the tankcircuit over a range of frequencies. The objective is to consume theleast amount of energy during the initial tuning process. Typically, the“tune” voltage level is 5% of the full scale voltage, where the fullscale voltage is the voltage at which the workpiece is intended to beheated.

After step 504, control passes to step 506. In step 506, RF power supply100 performs course tuning. That is, the RF power supply 100 determinesa course (i.e., rough estimate) of the tank circuit's resonantfrequency. The course estimate of the resonant frequency can bedetermined by sampling the tank circuit's admittance, impedance,reflected/forward power, etc . . . over a first predetermined frequencyrange. After step 506, control passes to step 508. In step 508, the RFpower supply 100 performs fine tuning. That is, the RF power supply 100determines a fine estimate (i.e., more precise estimate) of the tankcircuit's resonant frequency. The fine estimate can be determined bysampling the tank circuit's admittance, impedance, reflected/forwardpower, etc . . . over a second frequency range, which includes thecourse estimate of the resonant frequency. After step 508, controlpasses to steps 510 and 512 in parallel. In step 510, the RF powersupply 100 ramps (i.e., rapidly increases) the voltage output by the DCvoltage source 102 such that within approximately 30 milliseconds thevoltage increases from the “tuning” voltage level to approximately thefull scale voltage level. In step 512, the RF power supply 100continuously tracks the tank circuit's resonant frequency until a poweroff indication is received.

FIGS. 6A-7B further illustrate the process for inductively beating aworkpiece 109 where the tank circuit 108 is a parallel resonant tankcircuit 108 b, sensor 114 is an admittance sensor, such as the oneillustrated in FIG. 3A, and frequency controller 112 is implemented asshown in FIG. 2. It should be readily apparent to one skilled in therelevant art how to modify the processes illustrated in FIGS. 6A-7B fora series resonant tank circuit 108 c and for other types of sensors,such as an impedance sensor or a power sensor.

FIG. 6A further illustrates the course tuning process 506. The coursetuning process 506 begins at step 602 where main controller 10 signalsprocessor 202 to set the frequency of the alternating voltage producedby amplifier 104 to the upper frequency limit of the system. In oneembodiment of the present invention, the power supply has an upperfrequency limit of 485 KHz and a lower frequency limit of 50 KHz. Inanother embodiment, the upper frequency limit is 2 MHz and the lowerfrequency limit is 515 KHz. In still another embodiment the upperfrequency limit is 15 MHz and the lower frequency limit is 2 MHz. Theinvention, however, is by no means limited to these three embodiments.They are merely three examples of possible frequency ranges.

Next (step 604), after setting the frequency of the alternating voltageto the upper frequency limit, processor 202 reads the output of A/Dconverter 340 to determine an average admittance of tank circuit 108.The average admittance value determined by processor 202 is stored in afirst memory location or register within processor 202 (step 606). Thismemory location is referred to as “min-admittance.” The value of thefrequency at which tank circuit 108 is being driven at is stored in asecond memory location (step 608).

After step 608, control passes to step 610, where Processor 202 changesthe frequency (F) of the alternating voltage to a new frequencyaccording to the following formula: F=F(1−1I/Q), where Q is an estimateof the maximum quality factor (Q) for a given tank circuit 108. In thepreferred embodiment, it is assumed that Q is equal to 100.Consequently, the formula for calculating the new frequency is: F=(F)(0.99).

After changing the frequency, processor 202 reads the output of A/Dconverter 340 (step 612). The value read by processor 202 corresponds tothe admittance value of tank circuit 108 at the new frequency (F).Processor 202 then compares this admittance to the value stored inmin-admittance (step 614). If the admittance value is less than thevalue stored in min-admittance, then processor 202 stores the admittancevalue into min-admittance (step 618), thereby overwriting the value thatwas previously stored there, and stores the value of F into the secondmemory location (step 620). After step 620, control passes to step 616.Referring again to step 614, if the admittance value is greater than thevalue stored in min-admittance, then control immediately passes to step616.

In step 616, processor 202 determines if it has reached the lowerfrequency limit, (i.e., the processor 202 determines if F×0.99 isgreater than the lower frequency limit). If the lower frequency limithas not been reached, control passes back to step 610, otherwise thecourse tuning process is complete and control passes to step 508. At thecompletion of the course tuning process the value stored in the secondmemory location contains a course estimate of the resonant frequency oftank circuit 108. It should be noted that the course tuning processcould have begun at the lower frequency instead of at the upperfrequency limit.

FIG. 6B further illustrates the fine tuning process. The fine tuningprocess begins with step 632. In step 632, processor 202 sets thefrequency of the alternating voltage produced by amplifier 104 to afrequency (F) determined by the following formula: F=F_(c)×(1/0.99),where F_(c) is the course estimate of the resonant frequency. Next (step634), processor 202 reads the output of A/D converter 340 to determinean average admittance of tank circuit 108. The average admittance valuedetermined by processor 202 is stored in a first memory location (step636). The value of F, the frequency at which tank circuit 108 iscurrently being driven, is stored in a second memory location (step638).

After step 638, control passes to step 640 where Processor 202 changesthe frequency (F) of the alternating voltage to a new frequencyaccording to the following formula: F=F(1−1/(10×Q)), where Q is anestimate of the maximum quality factor (Q) for a given tank circuit 108.In the preferred embodiment, it is assumed that Q is equal to 100.Consequently, the formula for calculating the new frequency is:F=(F)(0.999).

After changing the frequency, processor 202 reads the output of A/Dconverter 340 (step 642). The value read by processor 202 corresponds tothe admittance value of tank circuit 108 at the new frequency (F).Processor 202 then compares this admittance to the value stored in thefirst memory location (step 644). If the admittance value is less thanthe value stored in min-admittance, then processor 202 stores theadmittance value into the first memory location (step 648), therebyoverwriting the value that was previously stored there, and stores thevalue of F into the second memory location (step 650). After step 650,control passes to step 646. Referring again to step 644, if theadmittance value is greater than the value stored in the first memorylocation, then control immediately passes to step 646.

In step 646, processor 202 determines if it has reached the lowerfrequency limit. The lower frequency limit is defined as F_(c)×0.99,where F_(c) is the course estimate of the resonant frequency. If thelower frequency limit has not been reached, control passes back to step640, otherwise the fine tuning process is complete and control passes tosteps 510 and 512 in parallel. At the completion of the fine tuningprocess the value stored in the second memory location contains the“fine” estimate of the resonant frequency of tank circuit 108.

FIG. 7A illustrates a resonant frequency tracking process 512 accordingto one embodiment. The process begins in step 702 where the processor202 sets the frequency of the alternating voltage provided to the tankcircuit 108 to the fine estimate of the resonant frequency determined instep 508. Next, the processor 202 measures the admittance, stores theadmittance value in a first memory location (Ar), and stores the currentfrequency in a second memory location (Fr) (step 704). The frequency ofthe alternating voltage is then reduced by an offset set amount to F−,where F−=Fr− offset. (step 706). The offset amount can be apredetermined constant or it can be a function of frequency. Preferably,the offset is the latter and is determined by the following formula:offset=Fr×0.1%. After the frequency is changed, the processor 202measures the admittance at the new frequency and stores the newadmittance value in a third memory location (A−) (step 708). Next (steps710 and 712), the frequency is changed to F+, where F+=Fr+ offset, and anew admittance reading is taken and stored in a fourth memory location(A+).

Processor 202 uses the values stored in A−, Ar, and A+ to determinewhether or not the present reference frequency, Fr, is the best estimateof the resonant frequency. If it is not the best estimate, processor 202sets Fr to either F+ or F− depending on the relative magnitudes of A−,Ar, and A+. It is assumed that if Ar<A− and Ar<=A+, then Fr is the bestapproximation of the resonant frequency. In this case, processor 202resets the frequency of the alternating voltage to Fr, measures theadmittance at Fr, and stores the admittance value in Ar. In the casewhere A−<=Ar and A−<=A+, processor 202 sets Fr to F− and sets Ar to A−.Finally, in the case where A+<Ar and A+<A−, processor 202 changes thefrequency of the alternating voltage to F+, sets Fr to F+, and sets Arto A+. The above is accomplished in steps 714-728. In step 730, theprocessor determines whether a fault has occurred or whether the heatshould be turned off. If either of those two conditions occur, controlpasses to step 732, where the process ends, otherwise control passesback to step 706.

FIG. 7B illustrates an alternative tracking process for tracking thetank circuit's resonant frequency. The process begins in step 751 wherethe RF power supply 100 sets the frequency (F) of the alternatingvoltage to the fine estimate of the resonant frequency determined instep 508. In step 752, “direction” is set to −1. In step 753, Fr is setto the fine estimate. In step 754, the admittance of the tank circuit ismeasured and the measured admittance value is stored in a first memorylocation. In step 756, the frequency (F) of the alternating voltage ischanged such that: F=Fr+(Fr)(0.1%)(direction). In step 758, theadmittance of the tank circuit is measured at the new frequency. In step760, it is determined whether the new measured admittance value is lessthan or equal to the admittance value stored in the first memorylocation. If it is, then control passes to step 762, otherwise controlpasses to step 764. In step 762, the new measured admittance value isstored in the first memory location and Fr is set to F. After step 762,control passes back to step 756. In step 764, the “direction” is changed(i.e., direction=(direction)(−1). In step 766, the frequency of thealternating voltage is set to Fr. After step 766, control passes back tostep 754. The process continues until a fault or heat off conditionoccurs.

As can be seen from the above described processes, the admittance oftank circuit 108 is continuously monitored throughout the entire heat onperiod. Based on the admittance of the tank circuit 108, the frequencyof the alternating voltage is adjusted so that the frequency of thealternating voltage matches the resonant frequency of the tank circuit108. In this manner, the desired amount of power is always delivered tothe load 109. It should be apparent, however, to one skilled in therelevant how to modify the above described processes for an RF powersupply that senses, among other things, impedance, forward power, andreflected power, as opposed to admittance.

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample, and not limitation. It will be understood by those skilled inthe art that various changes in form and detail may be made thereinwithout departing from the spirit and scope of the invention as definedby the following claims. Thus the breadth and scope of the presentinvention should not be limited by any of the above-described exemplaryembodiments, but should be defined only in accordance with the followingclaims and their equivalents.

What is claimed is:
 1. A radio frequency (RF) power system for heating aworkpiece, comprising: a circuit; a direct current (DC) voltage sourcethat provides a DC voltage within a first predetermined range; anamplifier, coupled to said DC voltage source, to provide an alternatingvoltage to said circuit; a frequency controller, coupled to saidamplifier, to control a frequency of said alternating voltage providedby said amplifier; and a power sensor being coupled to said circuit forproviding a signal to said frequency controller, wherein said frequencycontroller controls said frequency of said alternating voltage based onsaid signal provided from said power sensor, and wherein said powersensor senses both forward and reflected power, and said signalrepresents a ratio of said reflected power to said forward power,wherein said frequency controller comprises a resonant frequencytracking means for tracking a resonant frequency of said circuit whilethe workpiece is being heated, thereby enabling said frequencycontroller to control said frequency of said alternating voltage so thatsaid frequency of said alternating voltage tracks said resonantfrequency of said circuit while the workpiece is being heated, whereinsaid resonant frequency tracking means uses said signal that representsa ratio of said reflected power to said forward power to track saidresonant frequency of said circuit.
 2. A method for inductively heatinga workpiece, comprising the steps of: applying an alternating voltage toa circuit having a resonant frequency that changes while the workpieceis being heated, the alternating voltage having a frequency and having afirst voltage level; determining an estimate of the resonant frequencyof the circuit; setting said frequency of said alternating voltageprovided to said circuit to said determined estimate of said resonantfrequency; increasing said voltage level of the alternating voltage fromsaid first voltage level to a second voltage level after determiningsaid estimate of said resonant frequency of said circuit; and trackingsaid resonant frequency of said circuit until a heat off indication isgenerated.
 3. The method of claim 2, wherein the step of tracking saidresonant frequency of said circuit until a heat off indication isgenerated comprises the steps of: (a) determining a ratio of powerapplied to said circuit to power reflected from said circuit; (b)modifying said frequency of said alternating voltage applied to saidcircuit; (c) determining a ratio of power applied to said circuit topower reflected from said circuit; (d) comparing the magnitude of saidratio determined in step (a) to the magnitude of said ratio determinedin step (b); (e) modifying said frequency of said alternating voltageapplied to said circuit based at least in part on a result of saidcomparison of the magnitude of said ratio determined in step (a) to themagnitude of said ratio determined in step (b); and (f) repeating steps(a) through (e) until said heat off indication is generated.
 4. A radiofrequency (RF) power system for heating a workpiece, comprising: a tankcircuit; a direct current (DC) voltage source that provides a DC voltagewithin a first predetermined range; an amplifier, coupled to said DCvoltage source, to provide an alternating voltage to said tank circuit;a frequency controller, coupled to said amplifier, to control afrequency of said alternating voltage provided by said amplifier; and apower sensor being coupled to said tank circuit for providing a signalto said frequency controller, wherein said frequency controller controlssaid frequency of said alternating voltage based on said signal providedfrom said power sensor, and wherein said power sensor senses bothforward and reflected power, and said signal represents a ratio of saidforward power to said reflected power, wherein said frequency controllercomprises a resonant frequency tracking means for tracking a resonantfrequency of said tank circuit while the workpiece is being heated,thereby enabling said frequency controller to control said frequency ofsaid alternating voltage so that said frequency of said alternatingvoltage tracks said resonant frequency of said tank circuit while theworkpiece is being heated, wherein said resonant frequency trackingmeans uses said signal that represents a ratio of said forward power tosaid reflected power to track said resonant frequency of said tankcircuit.